The present invention relates to wireless communications, and in particular to a phased-array receiver adapted for use in wireless communication systems.
Omni-directional communication systems have been used extensively in various applications due, in part, to their insensitivity to orientation and location. Such systems, however, have a number of drawbacks. For example, the transmitter in such systems radiates electromagnetic power in all directions, only a small fraction of which reaches the intended receiver; this results in a considerable amount of waste in the transmitted power. Thus, for a given receiver sensitivity, a relatively higher electromagnetic power needs to be radiated by an omni-directional transmitter. Furthermore, because the electromagnetic propagation is carried out in all directions, the effects of phenomenon such as multi-path fading and interference are more pronounced.
In a single-directional communication system, power is only transmitted in one or more desirable directions. This is commonly achieved by using directional antennas (e.g., a parabolic dish) that provide antenna gain for some directions, and attenuations for others. Due to the passive nature of the antenna and the conservation of energy, the antenna gain and its directionality are related; a higher antenna gain corresponds to a narrower beam width and vice versa. Single-directional antennas are often used when the relative location and orientation of the transmitter and receiver are known in advance and do not change quickly or frequently. For example, this may be the case in fixed-point microwave links and satellite receivers. Additional antenna gain at the transmitter and/or receiver of such a communication system may improve the signal-to-noise-plus-interference ratio (SNIR), and thereby increase the effective channel capacity. However, a single-directional antenna is typically not well adapted for portable devices whose orientation may require fast and frequent changes via mechanical means.
Multiple antenna phased-array systems may be used to mimic a directional antenna with a bearing adapted to be electronically steered without requiring mechanical movement. Such electronic steering provides advantages associated with the antenna gain and directionality, while concurrently eliminating the need for frequent mechanical reorientation of the antenna. Moreover, the multiple antennas disposed in phased-array systems alleviate the performance requirements for the individual active devices disposed therein, and thus make these systems more immune to individual device failure.
Multiple antenna phased-array systems (hereinafter alternatively referred to as phased-arrays) are often used in communication systems and radars, such as multiple-input-multiple-out (MIMO) diversity transceivers and synthetic aperture radars (SAR). Phased arrays enable beam and null forming in various directions. However, conventional phased-arrays require a relatively large number of microwave modules, adding to their cost and complexity.
Higher frequencies offer more bandwidth, while reducing the required antenna size and spacing. The industrial, scientific, and medical (ISM) bands at 24 GHz, 60 GHz are suited for broadband communication using multiple antenna systems, such as phased-arrays, and the 77 GHz band is suited for automotive RADARS. Furthermore, the delay spread at such high frequency bands is smaller than those of lower frequency bands, such as 2.4 GHz and 5 GHz, thus rendering such high frequency bands more effective for indoor uses, allowing higher data rates. A ruling by the FCC has opened the 22-29 GHz band for automotive radar systems, such as autonomous cruise control, in addition to the already available bands at 77 GHz.
A phased-array includes a multitude of signal paths each connected to a different one of a multitude of receive antennas. The radiated signal is received at spatially-separated antenna elements (i.e., paths) at different times. A phased-array is adapted to compensate for the time difference associated with the receipt of the signals at the multitude of paths. The phased-array combines the time-compensated signals so as to enhance the reception from the desired direction(s), while concurrently rejecting emissions from other directions.
The antenna elements of a phased-array receiver may be arranged in a number of different spatial configurations. In the following, a brief description of a one-dimensional n-element linear array is provided with reference to FIG. 1. It is understood that similar descriptions also apply to the transmitters and are not discussed.
For a plane-wave, the signal arrives at each antenna element with a progressive time delay t at each antenna. This delay difference between two adjacent elements is related to their distance, d, and the signal angle of incidence with respect to the normal, θ, as follows:ct=d sin(θ)  (1)where c is the speed of light. In general, the signal arriving at the first antenna element is defined by:s0(t)=A(t)cos[wct+φ(t)]  (2)and where A(t) and φ(t) are the amplitude and phase of the signal and ωc is the carrier frequency. The signal received by the kth element may be expressed as:Sk(t)=S0(t−kτ)=A(t−kτ)cos[wct−kwcτ+φ(t−kτ)]  (3)
The equal spacing of the antenna elements is reflected in expression (3) as a progressive phase difference wcτ and a progressive time delay t in A(t) and φ(t). Adjustable time delay elements, τ′n (see FIG. 1) compensate for the signal delay and phase difference concurrently.
The combined signal Ssum(t) may be expressed as,
                                          S            sum                    ⁡                      (            t            )                          =                              ∑                          k              =              0                                      n              -              1                                ⁢                                    S              k                        ⁡                          (                              t                -                                  τ                  k                  ′                                            )                                                              =                              ∑                          k              =              0                                      n              -              1                                ⁢                                    A              ⁡                              (                                  t                  -                                      k                    ⁢                                                                                  ⁢                    τ                                    -                                      τ                    k                    ′                                                  )                                      ⁢                                                  ⁢                          cos              ⁡                              [                                                                            ω                      c                                        ⁢                    t                                    -                                                            ω                      c                                        ⁢                                          τ                      k                      ′                                                        -                                      k                    ⁢                                                                                  ⁢                                          ω                      c                                        ⁢                    τ                                    +                                      φ                    ⁡                                          (                                              t                        -                                                  k                          ⁢                                                                                                          ⁢                          τ                                                -                                                  τ                          k                          ′                                                                    )                                                                      ]                                                        
For τ′k =−kτ, the total output power signal is defined by:Ssum(t)=nA(t)cos[wct+φ(t)]
One known technique to obtain the time delay is by using broadband adjustable delay elements in the RF path. However, adjustable time delays at RF are challenging to integrate due to such non-ideal effects as, e.g., loss, noise, and nonlinearity.
While an ideal delay may compensate for differences in the arrival times at all frequencies, in narrowband applications it may be approximated differently. For a narrow band signal, A(t) and φ(t) change slowly relative to the carrier frequency, i.e., when τ<<τmodulate, the following approximations apply:A(t)≈A(t−τ)φ(t)=φ(t−kr)
Therefore, only the progressive phase difference wcτ requires compensation in expression (3). The time delay element may be replaced by a phase shifter which provides a phase-shift of θn to the nth path. To add the signal coherently, θn may be defined by:θn=nwct(8)
Unlike wideband signals, phase compensation for a narrowband signal may be made at various locations in the receiving chain, i.e., RF, LO, IF, analog baseband, or digital domain. An additional advantage of a phased-array is that it is adapted to attenuate the incident interference power from other directions. FIG. 2 shows the normalized array gain of the receive pattern of an 8-element array adapted for a narrowband signal having a 45° angel of incidences The antenna spacing is assumed to be equal to d=λ/2, as shown in FIG. 1, where λ is the wavelength. It is seen from FIG. 2 that the signals incident from other angles are suppressed. Furthermore, the signal power in each path of a phased-array may be weighted to adjust the null positions or to obtain a lower side-lobe level.
As is known, in a receiver, for a given modulation scheme, a maximum acceptable bit error rate (BER) is related to a minimum signal-to-noise ratio, SNR, at the baseband output of the receiver (input of the demodulator). For a given receiver sensitivity, the output SNR sets an upper limit on the noise figure of the receiver. The noise figure, NF, is defined as the ratio of the total output noise power to the output noise power caused only by the source. For a single path receiver, the following applies:10 Log(SNRout)=10 Log(SNRin)−NF
This expression, however, does not apply directly to a phased-array. FIG. 3 shows an n-element phased-array system 10, which is adapted to add input signals Sin. The noise received from each antenna is Nin Signal N1 in each element represents the noise introduced in the path. A different amplifier 12 disposed in each path, amplifies the received signal and delivers the amplified signal to combiner block 14. The output of combiner block 14 is supplied to amplifier 16 which also receives and amplifies noise N2. Output signal Sout generated by amplifier 16 is defined as below:Sout=n2G1G2Sin
Antenna's noise-contribution is, in part, determined by the temperature of the object(s) it is pointed at. When antenna noise sources are uncorrelated, the output total noise power is given by:Nout=n(Nin+N1)G1G2+N2G2
Thus compared to the output SNR of a single-path receiver, the output SNR of the array is improved by a factor between n and n2 depending on the noise and gain contribution of the different stages disposed in the array. The array noise factor may be defined as:
                    F        =                                                            n                ⁡                                  (                                                            N                      in                                        +                                          N                      1                                                        )                                            ⁢                              G                1                            ⁢                              G                2                                      +                                          N                2                            ⁢                              G                2                                                          n            ⁢                                                  ⁢                          N              in                        ⁢                          G              1                        ⁢                          G              2                                                              F        =                  n          ⁢                                          ⁢                                    SNR              in                                      SNR              out                                          Therefore, the SNR at the output of a phased-array may even be smaller than SNR at the input of the phased-array if n>F, where F is the noise factor. For a given NF, an n-path phased-array receiver has a sensitivity that is greater than that of a single-path phased-array by a factor of 10*log(n) in dB. For instance, the sensitivity of an 8-path phased-array receiver is 9 dB greater than that of a single-path phased-array.